There are some design complications encountered in applications that use single-supply circuitry to process an ac input signal. One drawback is that the signal often can’t be attenuated and level-shifted to fit within the supply range without unacceptably reducing the signal-to-noise ratio. This problem arises when attempting to detect the input signal crossings of both positive and negative voltage thresholds with good precision.
The circuit shown is well suited for such a task (see the figure). It was designed to extend the measurement range of a 4.5-digit, ±2-V digital panel meter to ±20 V, providing an auto-ranging capability. The output voltage allows easy interfacing with standard logic as well as analog switches. In addition, it controls the attenuation of the meter input and positions the display’s decimal point. While basically a Schmitt trigger, the circuit topology is unusual in that it features two independently settable thresholds for positive or negative signals.
The input signal, assumed to be low impedance, may be close to the positive threshold (i.e., one diode drop, VD, higher than the voltage at the noninverting input of amplifier A). In this case, the diode DN is cut off and diode DP conducts. Consequently, the positive threshold voltage, above which the output of A goes low, depends only on V+, R1, R2, and R3. Below this threshold, an interval of positive values of VIN may exist. Both diodes either conduct or are nearly off, depending on the voltage across R3 when DN is off. Yet this doesn’t induce an unwanted transition of the amplifier output. For negative values of VIN, DP is off and DN is on. The output of A goes low as the voltage at the noninverting input becomes less than the small positive voltage caused by the opamp bias current flowing through R4.
Neglecting the small voltage on R4, the threshold for negative input signals is given by the node equation: VNth + VD = −(V+ + VOUTR1/RH) (R2/R1). For positive input voltages, disregarding the loading effect of RH on the divider, the positive threshold is given by: VPth − VD = (V+ + VOUT R1/RH)(R2 + R3)/(R1 + R2 + R3). As a result, R1/RH expresses the ratio between the hysteresis and threshold voltages in both cases. Obviously, R1 is defined (and trimmed) first and the R3 span should guarantee the required control range for any value of R1.
This circuit is designed to give ±2-V thresholds with submillivolt hysteresis. It makes use of an OP184 operational amplifier, capable of withstanding inputs beyond its rails, with a 5-V supply. The currents flowing in the internal diodes (that protect the input transistors from differential overvoltage) can cause damage to the device. This also holds true for the currents moving through the internal junctions (when input voltage is beyond the rails).
To avoid this, series limiting resistors (RS) have been added to guarantee safe operation over the entire ±20-V input range. Differential input currents up to a few milliamps exist in normal operation. Therefore, it should be clear that the equations derived previously apply only to input voltages that are close to the threshold.
For higher-differential input voltages, the protection circuitry simply introduces attenuation in the first stage of the amplifier, with no effect on the comparator behavior. To deal with the nonnegligible common-mode currents, the resistance seen by the two inputs is balanced by proper selection of R4. This resistor provides an output path for bias current and properly biases the noninverting input when DP is off.