Electronic Design

Economical Flyback Converter Operates Off -48 V

Using a coupled inductor sharply cuts leakage between the input and output.

In telecom applications, the most common power-supply input voltage is −48 V. For years, telephone company central offices have supplied −48 V to operate standard phones, and they continue to do so. So for standardization purposes, the dc-dc converters supplying logic (+5 V) and analog (+12 V) voltages often use −48 V as the input supply.

The higher input voltage—higher than a +5-V supply, for example—has a substantially lower input current for a given output power. This significantly reduces wire losses and costs. Since phone lines span long distances, the power lost in the form of wiring losses can be significant.

To convert the higher and inverted input voltage of −48 V to lower, multiple positive voltages, flyback converters are often the best approach at low to medium powers. In a flyback converter, the output voltage can be configured to have any polarity and amplitude with respect to the input voltage. Also, multiple outputs are readily available just by adding secondary windings to the transformer.

The circuit shown in Figure 1 yields a low-cost solution for converting a −48-V supply to the dc voltages needed for a TV set-top box. The magnetic element is not a transformer but a coupled inductor, although additional windings can be added to it just the same. A transformer by definition does not store energy. Rather, it transforms one voltage or current level to another. For simplicity, the term "transformer" will be used to describe the multiwinding structure.

The coupled inductor is set up so the −48 V is impressed across the entire primary winding during the on-time of the primary switch. The secondary output voltages are tapped off the primary winding, and current is supplied to the output during the off-time.

As a result, the primary and secondary share the same winding by virtue of configuring the ground of the secondary as the positive terminal of the primary. This has the advantage of fewer windings over an all-isolated transformer structure. The transformer, then, is cheaper. Also, and perhaps more importantly, the leakage inductance between the input and output are drastically reduced. This reduction yields smaller voltage spikes on the primary switch at turn-off, providing less stringent snubbing requirements. Additionally, a lower-voltage MOSFET can be used, improving efficiency while reducing cost.

An isolated winding supplies the bias current for the SC1101 pulse-width-modulation (PWM) controller IC. This low-current winding needs to be isolated since its return is −48 V, not ground. In other words, the −48-V rail serves as the chip's ground.

When the main switch (Q2) is on, current flows in the primary winding for a time determined by the duty cycle and switching frequency. Current flows from ground, or the return of the secondary, to the −48-V rail through MOSFET Q2, storing energy in the transformer. Once the current ceases, the voltage at Q2's drain increases to meet the voltage required by the coupled inductor's outputs.

The voltage at pin 1 of the transformer is clamped to 10 V (plus a diode drop) during Q2's off-time (Fig. 1, again). The number of turns in the transformer is:

  • Pins 1-8 = 9 (10-V winding to Q2 drain
  • Pins 1-2 = 6 (6.5-V winding to 10-V winding)
  • Pins 2-6 = 9 (6.5-V winding to ground)
  • Pins 6-3 = 24 (−17-V winding to ground)
  • Pins 4-5 = 9 (bias winding to −48 V, isolated)

From this turns information, a primary versus secondary turns ratio (TR) can be derived. The derived TR is equivalent to having a standard transformer with separate primary and secondary windings. Remember that in this structure, the primary and secondary windings are the same. This reduces leakage, number of turns, and cost:

  • TR (10-V winding) = 9/(9 + 6 + 9) = 0.375
  • TR (6.5-V winding) = 6/(9 + 6 + 9) = 0.25
  • TR (−17-V winding) = 24/(9 + 6 + 9) = 1
  • TR (bias winding) = 9/(9 + 6 + 9) = 0.375

The duty cycle (D) of the flyback converter now can be calculated. Since the loop is closed around the highest power winding for best coupling, the 10-V, 1.5-A winding with a TR of 0.375 determines the duty cycle:

D = VOUT/(VOUT + TR × VIN)
= 10.6 V/(10.6 V + 0.375 × 48)
= 0.37

For the main output of 10 V, 0.6 V is added for the forward drop of the rectifier diode.

This is a suitable duty cycle since at high input voltages, input currents are lower, and narrow duty cycles do not result in excessive peak switch currents. Also, since the output ac ripple current is calculated using the off-time (1 − D), low values of D result in longer off-times and lower peak output currents, which reduces output-capacitor size and cost:

IOUT (avg) = IOUT (dc) = IOUT (pk) × (1 − D)

or:

IOUT (pk) = IOUT (avg)/(1 − D)

The nine additional turns between the drain of Q2 and the 10-V winding are used to optimize the duty cycle. If those additional turns were reduced to zero, D would be reduced to 0.22 at VIN = 48 V. This would yield too low a duty cycle at maximum input voltages of 60 V or higher, increasing the peak input currents.

When calculating the diodes' voltage requirements, the same voltage-divider scheme applies. When Q2 is conducting, the voltage is applied from ground to —48 V. The windings form a voltage divider, much the same as a resistance divider. With a maximum input voltage of −60 V, the voltage D5 must be able to block is equal to:

VD5 = 10.6 V − \[−60 × (9 + 6)/(9 + 9 + 6)\] = 48 V

and for D3:

VD3 = 7 V − \[−60 × (9/24)\] = 29.5 V

So, D3 can be a low-cost Schottky diode.

The converter achieves regulation by closing the feedback loop around the output with the highest power. With good transformer design, adequate coupling can be achieved on other outputs as well. If tighter regulation is desired, a linear regulator can be added to the outputs. The 10-V output is sensed through pnp transistor Q4, which acts as a level shifter. Q3, which can be a pnp transistor connected as a diode, compensates for Q4's VBE temperature coefficient, providing a temperature-compensated output. The present design can operate from −65°C to 80°C.

The current in R6 is equal to:

IR6 = (V10V/R6) + VBE − VBE = V10V/R6

This current is reflected in Q4's collector (ignoring negligible base current). R6 is selected for 1 mA of collector current, so the voltage at the feedback pin (FB) of the controller chip becomes:

VFB = (V10V/R6) × 1.24k

which equals the 1.25-V reference at the error-amplifier input when the desired regulation is achieved. Q4 must be rated to sustain a VCE equal to maximum input voltage plus the output voltage.

Figure 2 shows the waveforms at the transformer windings with respect to output ground. Using the turns ratio in a voltage-divider mode and per scope measurements of the 10-V winding, the voltage on Q2's drain flies up to:

VDRAIN × (T1-2 + T2-6)/(total turns) = 13 V

VDRAIN = 13 × \{24/(6 + 9)\} = 20.8 V

This agrees with the peak positive drain voltage on scope Ch 1. The drain goes to −48 V during Q2's on-time, since all the scope measurements are with respect to output ground and not the ground pin of the controller, which is −48 V.

Bootstrap And Startup: Since the −48-V supply provides the quiescent current for the controller IC and the current to drive the MOSFET gates, the bias current used by the chip at VCC of 5 to 6 V imposes an excessive power loss. The 15 to 20 mA of current used for bias at 5 V can add an additional power loss of:

PBIAS = (60 − 5) × 0.020 = 1.10 W

which is a significant percentage of the total output power. An additional winding can save this power by bootstrapping the bias supply once the PWM controller is on and the transformer voltages have been established. The cost of an additional winding is offset by the increase in efficiency.

The bootstrap operation works as follows. When power is first applied, Q1 turns on through the resistance divider R4 and R8, charging C3 through R5. Once the voltage on C3 exceeds 4.5 V, the PWM controller starts up, turning Q2 on and allowing current to flow through the transformer. Once the bootstrap power winding has enough voltage to turn on zener diode D6, Q5 will turn on. This action steals current away from the base at Q1 and turns Q1 off. The bias winding supplies a voltage that is approximately 6.5 V. Transistor Q1 must sustain the rail voltage of 60 V (max) and should be rated accordingly.

The SC1101, which can also be configured as a buck (step-down) controller, has the features to keep the flyback converter control loop simple. The dual-ground architecture allows for easy layout. The power ground (PGND) and the current-sense return CS(−) are referenced to the electrolytic input capacitor (C2) and Q2's source, while the GND pin is connected to the output dc return. The two grounds should be connected at the controller chip using a short connection.

The current-sense resistor (R13) limits the current in the primary on a pulse-by-pulse basis. It also acts as a soft-start resistor, charging the output capacitors and limiting the maximum current delivered to the secondary. This function prevents excessive current flow in the "coupled inductor" (the transformer), which could result in magnetic-core saturation.

There is no need for frequency compensation of the SC1101-based flyback converter, as long as some simple guidelines about the selection of external components and the parasitic parameters are observed. Flyback converters exhibit a loop phenomenon known as the right-half-plane (RHP) zero, which can lead to instabilities. In this circuit, the loop gain is designed to cross 0 dB at a frequency much lower than the RHP zero frequency. Typically, increasing the output capacitor size increases the margin for stability by lowering the LC pole frequency. The converter operates from an input of 30 to 60 V, with an efficiency of 92% at 60-V input and full load.

Finally, the transformer must be wound with judicious attention paid to leakage inductance to minimize voltage spikes on the MOSFET drain, as well as to improve coupling- and cross-regulation between windings. This reduces maximum voltage requirements for the MOSFET and snubbing requirements. Configuring the flyback converter's transformer as such by sharing the primary and secondary windings reduces the transformer's cost and complexity while improving coupling and regulation.

References
    1. "Worldwide Battery Market Status and Forecast," presented at the Power 2001 Conference by Hideo Takeshita, vice president of the Institute of Information Technology Ltd.; [email protected]

    2. For a description of Li-ion and Li-polymer charge/discharge methods and cell structures, see Sony's Li-ion Rechargeable Battery Catalog, p. 3, at www.sony.co.jp/en/Products/BAT/ION/index.html

    3. For some perspective on existing performance levels, read "Thinner Li-ion Batteries Power Next-Generation Portable Devices," Electronic Design, Feb. 7, 2000, p. 95-106; available online at www.elecdesign.com.



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