Latching Overvoltage Indicator Handles Large Overloads

June 4, 2001
This overvoltage detector can be connected to any dc power source (Fig. 1). Its purpose is to provide a visual indication of when the voltage exceeds a preset value, which may range from around 3 V...

This overvoltage detector can be connected to any dc power source (Fig. 1). Its purpose is to provide a visual indication of when the voltage exceeds a preset value, which may range from around 3 V to several hundred volts. The circuit also implements latching overvoltage detection, allowing it to capture transient overvoltage spikes as narrow as 30 µs.

Under "normal" voltage conditions, the whole circuit draws less than 25 µA, making it ideal for monitoring battery-powered systems. But despite its frugal diet, the detector is a fairly tough character, able to withstand continuous mains overloads without damage.

IC1, an LTC1541, is a micropower comparator, op amp, and voltage reference (1.20 V nominal) with a maximum quiescent current of just 13 µA. The op amp and the Q1/Q2 Darlington pair form a precision current sink. With R5 = 1.1 MΩ and R6 = 100k, the op amp puts a nominal 100 mV across R11. By doing so, it sinks 1 mA through the LED (D7). If the LED is a low-current type, such as the HLMP-D155, 1 mA is sufficient to achieve adequate brightness.

With no overvoltage condition present, IC1's comparator output is low. It clamps Q1's base to around 0.6 V via D6, which holds the current sink off. R1 and R2 determine the trip voltage, VT (the value of VIN at which the circuit indicates an overvoltage).

When the voltage (VR2) across R2 exceeds the 1.20-V reference, the comparator trips and its output goes high. D6 becomes reverse-biased, enabling the current sink to illuminate the LED. This provides immediate visual indication of the overvoltage event. At the same time, D5 becomes forward-biased, pulling the comparator's noninverting input toward VDD. The circuit is now latched, and the comparator output remains high even if VR2 subsequently falls below 1.20 V. By either disconnecting VIN or momentarily closing switch SW1, the circuit can be reset.

Under normal conditions, when D6 clamps Q1's base, the op-amp output rises toward VDD, desperately trying to bias Q1 on. Consequently, a substantial value is needed for R10 to minimize the op amp's output current (which adds to IC1's quiescent current). This means that only a few microamperes are available at Q1's base, dictating the use of the Darlington pair. The ZTX458 devices specified for Q1 and Q2 have an hfe high enough to furnish the LED with 1 mA when Q1's base current is as low as 1.5 µA. Since they're high-voltage types, rated to 400 V, they can withstand the 350-V peak voltage of the 240-VRMS mains supply.

If a MOSFET were substituted in place of the Darlington pair, its voltage and power ratings would have to cope with the worst-case anticipated overload. In addition, the MOSFET's gate-threshold voltage would need to be low enough to allow operation at lesser values of VIN.

D1 to D4 and R7 to R9 supply IC1 with overload protection. Diode D3 blocks any excessive negative inputs. The 1N4005 has a 600-V reverse voltage rating, which is more than adequate for a 240-VRMS overload. Diode D2 protects the comparator input against high positive voltages.

For precision voltage monitoring, R1 must be connected "upstream" of D3 as shown. Therefore, D1 is required to protect the comparator input against excessive negative voltages. Selecting a large value for R3 provides ample current limiting for D1 and D2. It also makes it necessary for the comparator to source negligible current through D5, permitting a relatively small value to be selected for R2 if required. Resistor R4 offers additional current limiting for the comparator input. Plus, it prevents the comparator output from being dragged low via D5 when SW1 is closed to reset the circuit.

Because the comparator's input bias current is very low (1 nA maximum), the voltage dropped across R3 and R4 is negligible. Zener diode D4 clamps IC1's positive supply to a safe value. Since the LTC1541's maximum working voltage is 12.6 V, using an 11-V zener is an appropriate selection.

A single resistor with suitable power and voltage ratings could be used instead of R7 to R9. But connecting three resistors from the MRS25 series (0.6-W, 250-V) in series guarantees that the parts will easily withstand a mains overload. The resistance values should be great enough to satisfy the 0.6-W power rating and ensure the zener's power dissipation is kept low. Using large values also minimizes the circuit's current draw when VIN exceeds the zener voltage. Yet the values shouldn't be too big, otherwise, IC1's VDD current will cause a relatively significant voltage drop. This would degrade (increase) the circuit's minimum working voltage.

When determining which values for R7 to R9 will be able to withstand mains overload, remember that D3 effectively half-wave rectifies the mains waveform. Therefore, even though the peak voltage at R7 will equal 1.414 × VRMS, the RMS voltage at this point will only be VRMS/2. For example, if VRMS is 240 V, each resistor will see just 40 VRMS. So selecting values of 3.6k will dissipate 0.44 W in each resistor. The resulting power dissipation in the 11-V zener would be 0.12 W.

With the circuit subjected to mains overload, the op amp's output briefly rises toward VDD on each positive cycle. It then settles down to the proper, lower value required to regulate the current sink. Ordinarily, this phenomenon would result in a large RMS voltage across R11, leading to an excessive LED current and excessive power dissipation in Q2. Fortunately, this problem is easily remedied by connecting Q3 across R11 as shown. Q3 normally has no effect on the current sink. But under mains overload conditions, Q3 keeps Q2's rms collector current below 2 mA. In doing so it ensures that the power dissipation is well within the ZTX458's 1-W limit.

The trip voltage, VT, is given by:

VT = VREF(R1+R2)/R2 (volts)

so: R1 = R2(VT − VREF)/VREF (ohms).

R1 and R2 should be as large as possible to minimize the current drawn from VIN and the power dissipation under overloads. The voltage ratings of R1, R2, and R3 must be able to withstand the maximum overvoltage at VIN.

A prototype circuit was built with R1 = 300k and R2 = 100k, equivalent to a nominal VT of 4.80 V. With VIN = 4.70 V (circuit untripped), the total current draw was just 20.6 µA. The circuit tripped when VIN exceeded 4.78 V. In the tripped state, the circuit's minimum working voltage (below which the LED current began to fall under 1 mA) was found to be VIN = 2.94 V.

The LTC1541 could be replaced by the pin-compatible MAX951, if required. Note, however, that the MAX951's maximum working voltage is only 7 V (9 V absolute maximum), requiring a lower zener voltage for D4.

If narrow transients on VIN cause nuisance tripping, a capacitor across SW1 will provide some immunity, but at the expense of response time.

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